Method for using a non-orthogonal pilot signal with data channel interference cancellation

ABSTRACT

A system and method for encoding/decoding data channels in a CDMA system having data channel interference cancellation, wherein data channel interference cancellation is used to remove unwanted non-orthogonal pilot signal components which are present within a demodulated data signal. This is accomplished by regenerating interference terms with respect to the non-orthogonal pilot signal and subtracting them from the demodulated data signal.

RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.11/178,767, filed Jul. 11, 2005 now U.S. Pat. No. 7,092,430, which is acontinuation of U.S. application Ser. No. 09/772,200 filed Jan. 29, 2001now U.S. Pat. No. 6,917,642, which claims the benefit of U.S.Provisional Application No. 60/184,365 filed on Feb. 23, 2000. Theentire teachings of the above applications are incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to the field of datacommunications and processing and, more particularly, to a method forencoding/decoding data channels in a CDMA system having data channelinterference cancellation.

2. Description of the Related Art

Code Division Multiple Access (CDMA) modulation is a multi-user accesstransmission scheme in which different users of the same transmissionmedium overlap both in frequency and in time. This is in contrast toFrequency Division Multiple Access (FDMA) in which users overlap intime, but are assigned unique frequencies, and Time Division MultipleAccess (TDMA) in which users overlap in frequency, but are assignedunique timeslots. According to CDMA, each user is assigned a unique codesequence that allows the user to spread its information over the entirechannel bandwidth, as opposed to particular sub-channel(s) in FDMA.Thus, signals from all users are transmitted over the entire channel. Toseparate out the signals for a particular user at a receiver, crosscorrelation is performed on the received signal using the same uniqueuser code sequence.

CDMA transmission is well known to those of skill in the art. Acomparison between CDMA and FDMA/TDMA may be found in Proakis, “DigitalCommunications”, Chapter 15, which is incorporated herein by reference.Also, an example of a combined approach for minimizing inter-userinterference (i.e., combining a Walsh basis within a group and aspreading sequence across groups) is the IS-95 system described inTIA/EIA/IS-95 “Mobile Station Compatibility Standard for Dual ModeWideband Spread Spectrum Cellular System”, which is incorporated hereinby reference.

An IS-95 CDMA system is unique in that its forward and reverse links(i.e., the base station to mobile station and mobile station to basestation) have different link structures. This is necessary toaccommodate the requirements of a land-mobile communication system. Theforward link consists of four types of logical channels, i.e., pilot,sync, paging, and traffic channels, with one pilot channel, one syncchannel, up to seven paging channels, and several traffic channels. Eachof these forward-linked channels is first spread orthogonally by itsWalsh function, and then spread by a pair of short PN sequences(so-called pseudonoise) each of which is a sequence of high data ratebits (“Chips”) ranging from −1 to +1 (polar) or 0 to 1 (non-polar).Subsequently, all channels in the system are added together to form thecomposite spread spectrum signal which is transmitted on the forwardlink.

The reverse link in the IS-95 CDMA system consists of two types oflogical channels, i.e., access and traffic channels. Each of thesereverse-link channels is spread orthogonally by a unique long PNsequence; hence each channel is recovered or decoded using the distinctlong PN code. In some instances, a pilot channel is not used on thereverse link based on the impracticality of each mobile stationbroadcasting its own pilot sequence. Additionally, the IS-95 CDMA systemuses 64 Walsh functions which are orthogonal to each other (i.e., theircross-product is equal to zero), and each of the logic channels on theforward link is identified by its assigned Walsh function. The Walshfunction is used to generate a code which is used to separate individualusers occupying the same RF band to avoid mutual interference on theforward link. The access channel is used by the mobile station tocommunicate with the base station when a traffic channel is not assignedto the mobile station. The mobile station uses the access channel togenerate call originations and respond to pages and orders. The basebanddata rate of the access channel is fixed at 4.8 kilobits per second(Kbps).

The pilot channel is identified by the Walsh function 0 (ω₀). Thischannel contains no baseband sequence information. The baseband sequenceis a stream of 0s which are spread by Walsh function 0, which is also asequence of all zeros. The resulting sequence (still all 0s) is thenspread or multiplied by a pair of quadrature PN sequences. Therefore,the pilot channel is effectively the PN sequence itself. The PN sequencewith a specified offset uniquely identifies the particular geographicalarea or sector from which the user is transmitting the pilot signal. Inan IS-95 CDMA system, both Walsh function 0 and the PN sequence operateat a rate of 1.2288 mega chips per second (Mcps). After PN spreading,baseband filters are used to shape the resultant digital pulses. Thesefilters effectively lowpass filter the digital pulse stream and controlthe baseband spectrum of the signal. As a result, the signal bandpossesses a sharper roll-off near the band edge. The pilot channel istransmitted continuously by the base station sector. The pilot channelprovides the mobile station with timing and phase reference. Themeasurement of the signal-to-noise ratio of the pilot channel by themobile station also provides an indication of the strongest servingsector of that mobile. Here, the signal-to noise is the energy per chipper interference density, or E_(c)/I₀ where E_(c) is the energy per chipand I₀ is the interference density.

Unlike the pilot channel, the sync channel carries baseband information.The baseband information is contained in the sync channel message whichnotifies the mobile of information concerning system synchronization andparameters. Similar to the sync channel, the paging channel also carriesbaseband information. However, unlike the sync channel, the pagingchannel transmits at a higher rate, i.e., at either 4.8 or 9.6 Kbps.

The forward and reverse traffic channels are used to transmit user dataand voice; signaling messages are also sent over the traffic channel.The structure of the forward traffic channel is similar to that of thepaging channel, while the structure of the reverse traffic channel issimilar to that of the access channel. The only difference is that theforward traffic channel contains multiplexed power control bits (PCBs)and the reverse traffic channel contains a data burst randomizer whichis used to generate a masking pattern of 0s and 1s to randomly maskredundant data.

The techniques for separating signals in time (i.e., TDMA), or infrequency (i.e., FDMA) are relatively simple ways of ensuring that thesignals are orthogonal and noninterfereing. However, in CDMA, differentusers occupy the same bandwidth at the same, but are separated from eachother via the use of a set of orthogonal waveforms, sequences, or codes.Two real-valued waveforms x and y are said to be orthogonal if theircross correlation R_(xy) over time period T is zero, where

$\begin{matrix}{{R_{xy}(0)} = {\int_{0}^{T}{{x(t)}{y(t)}\ {\mathbb{d}t}}}} & \left( {{Eq}.\mspace{14mu} 1} \right)\end{matrix}$In discrete time, the two sequences x and y are orthogonal if theircross-product R_(xy)(0) is zero. The cross product is defined as

$\begin{matrix}{{{R_{xy}(0)} = {{x^{T}y^{T}} = {\sum\limits_{i = 1}^{I}{x_{i}y_{i}}}}}{where}{x^{T} = \left\lbrack {x_{1}\mspace{14mu} x_{2}\cdots\mspace{11mu} x_{i}} \right\rbrack}{y^{T} = \left\lbrack {y_{1}\mspace{14mu} y_{2}\cdots\mspace{11mu} y_{i}} \right\rbrack}} & \left( {{Eq}.\mspace{14mu} 2} \right)\end{matrix}$In this case, T denotes the vector transpose, i.e., a column representedas a row or vice versa. For example, the following two sequences orcodes, x and y are orthogonal:x ^(T)=[−1−111]y ^(T)=[−111−1]because their cross-correlation is zero; that isR _(xy)(0)=x ^(T) y ^(T)=(−1)(−1)+(−1)(1)+(1)(1)+(1)(−1)  (Eq. 3)In order for the set of codes to be used in a multiple access scheme,additional properties are required. That is, in addition to the zerocross-correlation property, each code in the set of orthogonal codesmust have an equal number of 1s and −1s. This property provides eachparticular code with the required pseudorandom characteristic. Anadditional property is that the dot product of each code scaled by theorder of the code must equal to 1. The order of the code is effectivelythe length of the code, and the dot product is defined as a scalarobtained by multiplying the sequence by itself and summing theindividual terms. This is given by the following relationship:

$\begin{matrix}{{R_{xx}(0)} = {{x^{T}x} = {\sum\limits_{i = 1}^{I}{x_{i}x_{i}}}}} & \left( {{Eq}.\mspace{14mu} 4} \right)\end{matrix}$

The increasing use of wireless telephones and personal computers has ledto a corresponding demand for such advanced telecommunicationstechniques as CDMA, FDMA and TDMA, which were once thought to be onlymeant for use in specialized applications. In the 1980's wireless voicecommunication became widely available through the cellular telephonenetwork. Such services were at first typically considered to be theexclusive province of the businessman because of high subscriber costs.The same was also true for access to remotely distributed computernetworks, whereby until very recently, only business people and largeinstitutions could afford the necessary computers and wireline accessequipment. As a result of the widespread availability of bothtechnologies, the general population now increasingly wishes to not onlyhave access to networks such as the Internet and private intranets, butalso to access such networks in a wireless manner as well. This is ofparticular concern to the users of portable computers, laptop computers,hand-held personal digital assistants and the like who prefer to accesssuch networks without being tethered to a telephone line.

However, there is still no widely available satisfactory solution forproviding low cost, broad geographical coverage, high speed access tothe Internet, private intranets, and other networks using the existingwireless infrastructure. This situation is a result of several factors.For one, the typical manner of providing high speed data service in thebusiness environment over the wireline network is not readily adaptableto the voice grade service which is available in most homes or offices.Additionally, such standard high speed data services do not lendthemselves well to efficient transmission over standard cellularwireless handsets. Furthermore, the existing cellular network wasoriginally designed only to deliver voice services. As a result, theemphasis in present day digital wireless communication schemes lies withvoice, although certain schemes such as CDMA do provide some measure ofasymmetrical behavior for the accommodation of data transmission. Forexample, the data rate on an IS-95 forward traffic channel can beadjusted in increments from 1.2 Kbps to up to 9.6 Kbps for so-calledRate Set 1, and for increments from 1.8 Kbps up to 14.4 Kbps for RateSet 2.

Existing systems therefore typically provide a radio channel which canaccommodate maximum data rates only in the range of 14.4 Kbps at best inthe forward direction. Such a low rate data channel does not directlylend itself to transmitting data at rates of 28.8 or even 56.6 Kbpswhich are now commonly available with conventional modem type equipment.Data rates at these levels are rapidly becoming the minimum acceptablerates for activities such as Internet access. Other types of datanetworks using higher speed building blocks such as Digital SubscriberLine (xDSL) service are just now coming into use. However, the cost ofxDSL service has only recently been reduced to the point where it isattractive to the residential customer.

Although xDSL and Integrate Services Digital Network (ISDN) networkswere known at the time that cellular systems were originally deployed,for the most part, there is no provision for providing higher speed ISDNor xDSL grade data services over cellular networks. Unfortunately, inwireless environments, access to channels by multiple subscribers isexpensive and there is competition for them. Whether the multiple accessis provided by the traditional FDMA using analog modulation on a groupof radio carriers, or by the newer digital modulation schemes whichpermit sharing of a radio carrier using TDMA or CDMA, the nature of theradio spectrum is that it is a medium which is expected to be shared.This is quite different from the traditional environment for datatransmission, in which the wireline medium is relatively inexpensive toobtain, and is therefore not typically intended to be shared.Accordingly, it is apparent that there is a need to provide a systemwhich supports higher speed ISDN or xDSL grade data services overcellular network topologies. In particular, what is needed is anefficient scheme for supporting wireless data communication such as fromportable computers to computer networks such as the Internet and privateintranets using widely available infrastructure.

Most modem wireless standards in widespread use such as CDMA do notprovide an adequate structure with which to support the most commonactivities, such as web page browsing. In the forward and reverse linkdirection, the maximum available channel bandwidth in an IS-95 type CDMAsystem is only 14.4 Kbps. Due to IS-95 being circuit-switched, there areonly a maximum of 64 circuit-switched users that can be active at onetime. In practicality, this limit is difficult to attain, and 20 or 30simultaneous users are typically active at one time. Furthermore,existing CDMA systems require certain operations before a channel can beused. For example, both access and traffic channels are modulated byso-called long code pseudonoise (PN) sequences. In addition, in orderfor the receiver to work properly it must first be synchronized with thetransmitter. The setting up and tearing down of user channels thereforerequires overhead to perform such synchronization. This overhead resultsin a reduction of the system data rate which produces a noticeable delayto a user of a subscriber unit. Moreover, in the presence of benign cellconditions, the data rate of a conventional CDMA system may becomelimited by the number of available orthogonal code channels.

SUMMARY OF THE INVENTION

The present invention is directed to a method for encoding/decoding datachannels in a system having data channel interference cancellation. Inaccordance with the invention, the data rate of a system for a givenuser is increased by using a non-orthogonal pilot signal forchannelization. As a result, one or more orthogonal channels becomeavailable for user traffic, rather than for use by the pilot channel.This leads to a reduction in the number of occupied orthogonal channelsand an increase in system capacity available for each user due to theattainment of higher data rates which permit faster data delivery tosystem subscribers.

The use of a non-orthogonal pilot signal requires interferencecancellation to remove the modulation effects of the pilot signal uponthe data signal. This is accomplished by regenerating interference termswith respect to the non-orthogonal pilot signal and subtracting themfrom the demodulated data signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be more readily understood by one skilled inthe art with reference being had to the following detailed descriptionof the preferred embodiments thereof, taken in conjunction With theaccompanying drawings wherein like elements are designated by identicalreference numerals throughout the several views, and in which:

FIG. 1 is a block diagram of a wireless communication system which usesinterference cancellation on the pilot channel in accordance with theinvention;

FIG. 2 is a schematic block diagram of a CDMA transceiver forimplementing the method in accordance with the present invention;

FIG. 3 is an illustration of a pilot/data spreader of FIG. 2;

FIG. 4 is an illustration of a data despreader of FIG. 2;

FIG. 5 is an illustration of a pilot despreader of FIG. 2;

FIG. 6 is an illustration of an interference cancellor of FIG. 2;

FIG. 7 is an illustration of a dot product calculator of FIG. 2; and

FIGS. 8A and 8B are flow charts illustrating the steps of the methodaccording to the invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a block diagram of a wireless communication system 100 whichuses data channel interference cancellation to remove unwantednon-orthogonal pilot signal components present within the data signal.This results in a reduction in the number of occupied orthogonalchannels and an increase in system capacity. This yields an increase inthe system data rate which results in a reduction of the delayexperienced by the user of the subscriber unit. As a result, high speeddata and voice service over a wireless connection is achieved.

The system 100 includes two different types of components, such assubscriber units 101-1, 101-2, . . . , 101-u (collectively, thesubscriber unit 101) and one or more base stations 170. The subscriberunits 101 and base stations 170 cooperate to provide the functionsnecessary to provide wireless data services to a portable computingdevice 110 such as a laptop computer, portable computer, personaldigital assistance (PDA) or the like associated with a subscriber unit101. The base station 170 also cooperates with the subscriber units 101to permit the ultimate transmission of data to and from the subscriberunit 101 and the public switch telephone network (PSTN) 180. Moreparticularly, data and/or voice services are also provided by thesubscriber unit 101 to the portable computer 110 as well as one or moredevices such as telephones. The telephones themselves may in turn beconnected to other modems and computers which are not shown in FIG. 1.

The subscriber unit 101 itself may include a modem, such as an ISDNmodem 120, a device referred to herein as a protocol converter 130 whichperforms various functions including spooling 132 and bandwidthmanagement 134, CDMA transceiver 140, and subscriber unit antenna 150.The various components of the subscriber unit 101 may be realized indiscrete devices or as an integrated unit. For example, an existingconventional ISDN modem 120 such as is readily available from any numberof manufacturers may be used together with existing CDMA transceivers140. In this case, the necessary additional functions may be providedentirely by the protocol converter 130 which may be sold as a separatedevice. Alternatively, the ISDN modem 120, protocol converter 130 andCDMA transceiver 140 may be integrated as a complete unit and sold as asingle subscriber unit device 101. Other types of interface connectionssuch as Ethernet or PCMCIA may be used to connect the computing deviceto the protocol converter 130. The device may also interface to anEthernet interface rather than an ISDN “U” interface.

The ISDN modem 120 converts data and voice signals between the formatused by the terminal equipment 110 and the format required by thestandard ISDN “U” interface. The U interface is a reference point inISDN systems that designates a point of the connection between thenetwork termination (NT) and the telephone company.

The protocol converter 130 performs spooling 132 and basic bandwidthmanagement 134 functions. In general, spooling 132 consists of insuringthat the subscriber unit 101 communicates with the terminal equipment110 which is connected to the public switched telephone network 180 onthe other side of the base station 170 at all times. The bandwidthmanagement function 134 is responsible for allocating and deallocatingCDMA radio channels 160 as required. Bandwidth management 134 alsoincludes the dynamic management of the bandwidth allocated to a givensession by dynamically assigning sub-portions of the CDMA radio channels160. The CDMA transceiver 140 accepts the data from the protocolconverter 130 and reformats the data into the appropriate form fortransmission through the subscriber unit antenna 150 over CDMA radiolink 160-1. The CDMA transceiver 140 may operate over only a single 1.25MHz radio frequency channel, or may be tunable over multiple allocatableradio frequency channels.

CDMA signal transmissions from the subscriber units 101 are received andprocessed by the base station equipment 170. The base station equipment170 typically includes multichannel antennas 171, multiple CDMAtransceivers 172 and a bandwidth management function 174. Bandwidthmanagement 174 controls the allocation of CDMA radio channels 160 andsubchannels, in a manner analogous to the subscriber unit 101.Transceiver 172 demodulates the received CDMA signals, and the basestation 170 couples the demodulated radio signals to the PSTN 180 in amanner which is well known in the art. For example, the base station 170may communicate with the PSTN 180 over any number of different efficientcommunication protocols such as primary rate ISDN, or other LAPD basedprotocol such as IS-634 or V5.2.

It should also be understood that data signals travel bidirectionallyacross the CDMA radio channels 160. In other words, data signalsreceived from the PSTN 180 are coupled to the portable computer 110 in aforward link direction, and data signals originating at the portablecomputer 110 are coupled to the PSTN 180 in a reverse link direction.

Each of the CDMA transceivers such as transceiver 140 in the subscriberunit 101, and transceivers 172 in the base station 170, are capable ofbeing tuned at any given point in time to a given 1.25 Megahertz radiofrequency channel. It is generally understood that such 1.25 MHz radiofrequency carrier provides, at best, a total equivalent of about 500,600 kbps maximum data rate transmission within acceptable bit error ratelimitations.

FIG. 2 is a schematic block diagram of CDMA transceivers 140, 172 of thewireless communication system 100 for implementing the method accordingto the present invention. Specifically, FIG. 2 is a block diagram of atransmitter portion of a transceiver 140 and a receiver portion oftransceiver 172. Initially, pilot spreader 201 is used to modulate anon-orthogonal pilot signal such that the pilot signal is spread over anentire channel bandwidth. Concurrently, data spreader 204 is used tospread data over the same channel bandwidth. The spread pilot and datasignals are then combined to form a composite signal S(t) which istransmitted to base station 170 for despreading by pilot despreader 202and data despreader 205, respectively. The despreaders 202, 205 are usedto recover the non-orthogonal pilot signal and the data signal,respectively, from the transmitted composite signal S(t). The outputs ofthe pilot despreader 202 and data despreader 205 are fed to aninterference canceller 203 which is used to remove interferenceintroduced into the data signal by the non-orthogonal pilot signal. Oncethe interference from the non-orthogonal pilot signal is removed by theinterference canceller 203, the original data is recovered via dotproduct calculator 206 and output for later processing by acommunications system (not shown).

FIG. 3 is a block diagram of a pilot/data spreader 201 and 204 of FIG. 2which are used to modulate the non-orthogonal pilot and data signalssuch that they are spread over an entire channel bandwidth. At nodes 201a and 201 b of the pilot spreader 201, a non-orthogonal pilot signal Pis modulated by a channel code p_(c), which is used to uniquely identifythe transmitted pilot signal P. At nodes 204 a and 204 b of the dataspreader 204, a data signal which is split into sub-band data I and Q ismixed with a signal g_(i) which represents a specific channel code of auser (I and Q represent the in-phase and quadrature portions of the datasignal, respectively). At node 201 c/204 c, the output signal from node201 a is summed with the output signal from node 204 a to produce aresultant signal. Simultaneously, at node 201 d/204 d, the output signalfrom node 204 b is summed with the output signal from node 201 b toproduce a resultant signal.

At nodes 201 e/204 e and 201 f/204 f, the resultant signals are eachmodulated by a PN code α. Next, in order to provide baseband or phasediscrimination between the I and Q sub-band portions of the data signal,the output signals of nodes 201 e/204 e and 201 f/204 f are modulated(i.e., spread) by channel separation signals w_(I) and s_(Q),respectively, at nodes 201 g/204 g and 201 h/204 h, respectively. Inthis case, the channel separation signals w_(I) and w_(Q) belong to afamily of orthogonal functions such as those disclosed in U.S. Pat. No.4,460,992 to Gutleber, which is incorporated herein by reference as ifset forth expressly. Each respective channel separation signal spreadsthe in-phase portion and quadrature portion of the data signal toproduce composite signals. At the nodes 201 i/204 i and 201 j/204 j, therespective composite output signals from nodes 201 g/204 g and 201 h/204h are subsequently modulated by respective cosine and sine functions(i.e., cos(wt+θ) and sin(wt+θ)). The output signals from nodes 201 i/204i and 204 j/201 j are then summed to form a composite signal S(t) givenby the following relationship:S(t)=Pap_(c) w _(I) cos(wt+θ)+Pap_(c) w _(Q) sin(wt+θ)+I _(n) aw _(I) g_(i) cos(wt+θ)+Q _(n) αw _(Q) g _(i) sin(wt+θ)   (Eq. 5)

The signal given by the relationship in equation 5 is transmitted tobase station 170 which contains a data despreader 205 (see FIG. 6) foruse in the demodulation of the transmitted composite signal S(t) torecover the original data signal.

FIG. 4 is a schematic block diagram of a data despreader 205 which isused in the recovery of the originally transmitted data signal. In thedata despreader 205 shown in FIG. 4, the signal S(t) given in equation 5is initially decoded by demodulating S(t) by cos(wt) and sin(wt) atnodes 205 a and 205 b, respectively to produce resultant output signals.Next, at nodes 205 c and 205 d, the resultant output signals from nodes205 a and 205 b are demodulated by the PN code α. The output signals ofnodes 205 c and 205 d are each demodulated by the channel separationfunction w_(Q) at nodes 205 f and 205 g, respectively. Concurrently, theoutput signal of node 205 c is demodulated by the channel separationfunction w, at node 205 e, while at node 205 h the output signal of node205 d is demodulated by a channel separation function −w_(I) which is acomplex conjugate of the channel separation function w_(I). The outputsignals of nodes 205 e, 205 f, 205 g and 205 h are respectivelydemodulated at nodes 205 i, 205 j, 205 k and 205 l by the channel codeof a user g_(i).

Given two codes A and B of length n, an integration and dump functionoccurs when the lengths of the codes are matched, multiplied together,integrated and the result output for further processing. In this manner,an integration and dump function is then performed at nodes 205 m-205 p,respectively, upon the output signals of nodes 205 i-205 l to obtain thefollowing relationships:

$\begin{matrix}{{\sum\limits_{N}{\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{Q}g_{i}}} = {{\frac{N}{2}Q_{n}{\sin(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}{\sin(\theta)}}}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}w_{I}w_{Q}{\cos(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 6} \right) \\{{\sum\limits_{N}{\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{I}g_{i}}} = {{\frac{N}{2}I_{n}{\cos(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}{\cos(\theta)}}}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}w_{I}w_{Q}{\sin(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 7} \right) \\{{\sum\limits_{N}{\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){a\left( {- w_{i}} \right)}g_{i}}} = {{\frac{N}{2}I_{n}{\sin(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}{\sin(\theta)}}}} - {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}w_{I}w_{Q}{\cos(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 8} \right) \\{{\sum\limits_{N}{\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){aw}_{Q}g_{i}}} = {{\frac{N}{2}Q_{n}{\cos(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}{\cos(\theta)}}}} - {\frac{1}{2}{\sum\limits_{N}{{Pg}_{i}p_{c}w_{I}w_{Q}{\sin(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 9} \right)\end{matrix}$where each summation term in equations 6-9 represents interference dueto the pilot signal which must be removed to accurately reconstruct theoriginally transmitted data signal, and each N in the summation is theprocessing gain.

FIG. 5 is an illustration of the pilot despreader 202 which is used torecover the originally transmitted pilot signal P. To accomplish this,the transmitted composite signal S(t), given by the relationship inequation 14, is demodulated by cosine and sine functions (i.e., cos(ωt)and sin(ωt)) at nodes 202 a and 202 b. Next, the output signals fromnodes 202 a and 202 b are demodulated by the PN code a at nodes 202 cand 202 d, respectively. The output signals from nodes 202 c and 202 dare each demodulated by the channel separation function w_(Q) at nodes202 f and 202 g, respectively. Concurrently, the output signal of node205 c is demodulated by the channel separation function w_(I), at node202 e, while at node 202 h the output signal of node 202 d isdemodulated by the channel separation function −w_(I). The outputsignals of nodes 202 e, 202 f, 202 g and 202 h are respectivelydemodulated at nodes 202 i, 202 j, 202 k and 202 l by the channel codep_(c) which is used to uniquely identify the transmitted pilot signal P.After demodulating the output signals of nodes 202 i-202 l, theintegration and dump function is performed to obtain the output signalsgiven by the following relationships at nodes 205 m, 205 n, 205 o and205 p, respectively

$\begin{matrix}{{\sum\limits_{N}{\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{I}p_{c}}} = {{\frac{N}{2}P\;{\cos(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{I_{n}g_{i}p_{c}\cos\;\theta}}} + {\frac{1}{2}{\sum\limits_{N}{Q_{n}g_{i}p_{c}w_{i}w_{Q}{\sin(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 10} \right) \\{{\sum\limits_{N}{\left( {{P_{I}(t)} + {S_{I}(t)}} \right){aw}_{Q}p_{c}}} = {{\frac{N}{2}P\mspace{11mu}{\sin(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{Q_{n}g_{i}p_{c}\sin\;\theta}}} + {\frac{1}{2}{\sum\limits_{N}{I_{n}g_{i}p_{c}w_{i}w_{Q}{\cos(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 11} \right) \\{{\sum\limits_{N}{\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){a\left( {- w_{I}} \right)}p_{c}}} = {{\frac{N}{2}P\mspace{11mu}{\sin(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{I_{n}g_{i}p_{c}\sin\;\theta}}} - {\frac{1}{2}{\sum\limits_{N}{Q_{n}g_{i}p_{c}w_{i}w_{Q}{\cos(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 12} \right) \\{{\sum\limits_{N}{\left( {{P_{Q}(t)} + {S_{Q}(t)}} \right){aw}_{Q}p_{c}}} = {{\frac{N}{2}P\;{\cos(\theta)}} + {\frac{1}{2}{\sum\limits_{N}{Q_{n}g_{i}p_{c}\cos\;\theta}}} - {\frac{1}{2}{\sum\limits_{N}{I_{n}g_{i}p_{c}w_{i}w_{Q}{\sin(\theta)}}}}}} & \left( {{Eq}.\mspace{14mu} 13} \right)\end{matrix}$

where N is the processing gain.

As shown in FIG. 5, four output signals are generated which each containinterference as a result of the demodulation process. Equations 10-13represent the output signal at nodes 202 m, 202 n, 202 o and 202 p,respectively. In this case, the eight summation terms in equations 10-13represent the interference added to the pilot signal as a result of thedemodulation process. At node 202 q, the output signals from nodes 202 mand 202 o are subsequently subjected to an additional integration anddump function, while the integration and dump function is performed onthe output signals from nodes 202 n and 202 p at node 202 r. As aresult, the signals are filtered such that the interference is removedand the originally transmitted pilot signal P is recovered.

Along with the output of the pilot despreader 202, the output of thedata despreader 205 is provided to an interference canceller 203 shownin FIG. 6. The output of the pilot despreader is fed to the input of theinterference canceller 203, and the output of the interference canceller203 is subtracted from the output of the data despreader 205 in a mannerwhich is known to yield I and Q sub-band data signals which do notcontain interference associated with the pilot signal P.

The interference canceller 203 shown in FIG. 6 is used to remove theinterference associated with the pilot signal P which is introduced intothe data signal during the demodulation process. The interference addedto the data signal is represented by the summation terms in therelationships given in equations 6-9. To remove the interference fromthe despread data signals, the P cos(θ) and P sin(θ) inputs of theinterference canceller 203 are each modulated by the channel code p_(c)at nodes 203 a and 203 b, respectively. Next, the output signals ofnodes 203 a and 203 b are each modulated by the group user channel codeg_(i). At this point, an integration of the output signals of nodes 203c and 203 d is performed to yield respective first and secondinterference terms given by the following relationships:

$\begin{matrix}{{\frac{P\;{\cos(\theta)}}{2}{\sum\limits_{N}{g_{i}p_{c}}}}{and}} & \left( {{Eq}.\mspace{14mu} 14} \right) \\{\frac{P\;{\sin(\theta)}}{2}{\sum\limits_{N}{g_{i}p_{c}}}} & \left( {{Eq}.\mspace{14mu} 15} \right)\end{matrix}$

where N is the processing gain.

Next, the output signals from nodes 203 c and 203 d are modulated by thew, channel separation function at nodes 203 g and 203 h, respectively.The output signals from nodes 203 g and 203 h are then modulated by thechannel separation function w_(Q) at nodes 203 i and 203 j,respectively. An integration of the output signals from nodes 203 i and203 j is performed at nodes 203 k and 203 l to yield respective thirdand fourth interference terms given by the following relationships:

$\begin{matrix}{{\frac{P\;{\cos(\theta)}}{2}{\sum\limits_{N}{g_{i}p_{c}w_{I}w_{Q}}}}{and}} & \left( {{Eq}.\mspace{14mu} 16} \right) \\{\frac{P\;{\sin(\theta)}}{2}{\sum\limits_{N}{g_{i}p_{c}w_{I}w_{Q}}}} & \left( {{Eq}.\mspace{14mu} 17} \right)\end{matrix}$

where N is the processing gain.

The relationships expressed in equations 14-17 are subtracted from therespective expressions found in equations 6-9 to remove the interferencefrom the I and Q sub-band data signals. At this point, once theinterference is removed from the data signal, complete recovery of thedata signal is possible.

FIG. 7 is an illustration of an exemplary dot product calculator 206 forperforming a dot product calculation to recover the original datasignal. After removal of the interference terms given in equations14-17, each respective portion of the I_(n) and Q_(n) sub-band datasignals is forwarded to the dot product calculator 206. The respectivecosine and sine portions of the pilot signal P which are output from thepilot despreader 202 are also forwarded to the dot product calculator206, as shown in FIG. 7. At nodes 206 a and 206 b, the cosine portion ofthe pilot signal P is modulated by the cosine portions of the I_(n) andQ_(n) sub-band data signals. Simultaneously, at nodes 206 c and 206 d,the sine portion of the pilot signal P is modulated by the sine portionsof the I_(n), and Q_(n) sub-band data signals. At node 206 e, the outputsignal of nodes 206 a and 206 c are summed together to yield an outputsignal given by the following relationship:PI _(n) cos(θ−{circumflex over (θ)})≈PI _(n)  (Eq. 18)At node 206 f, the output signals of nodes 206 b and 206 d are summedtogether to yield another output signal given by the followingrelationship:PQ _(n) cos(θ−{circumflex over (θ)})≈PQ _(n)  (Eq. 19)where the ^ term in equations 18 and 19 indicates a coarse estimate ofthe phase over one symbol (i.e., the number of chips per signal). Atthis point, one skilled in the art will readily appreciate thatequations 18 and 19 represent the originally transmitted I and Qsub-band data signals, where each sub-band is multiplied by the pilotsignal P.

FIGS. 8A and 8B are flow charts of the method for using a non-orthogonalpilot signal according to the invention. In step 10, a non-orthogonalpilot signal P is modulated by a channel code p_(c). Simultaneously, adata signal which is split into sub-band data I and Q is mixed with aspecific channel code of a user g_(i). In step 20, the non-orthogonalpilot signal is then summed with the I and Q sub-band data signals toproduce resultant signals.

In step 30, the resultant signals are then modulated by a PN code α. Instep 40, to provide baseband or phase discrimination between the I and Qsub-band portions of the data signal, the resultant output signals aremodulated (i.e., spread) by channel separation signals w_(I), and w_(Q).In step 50, the respective composite output signals are modulated byrespective cosine and sine functions (i.e., cos(wt+θ) and sin(wt+θ). Instep 60, the cosine and sine output signals are then summed to form thecomposite signal S(t) which is transmitted to the base station 170.

In step 70, the composite signal S(t) is initially decoded bydemodulating it with cos(wt) and sin(wt). Next in step 80, the resultantoutput signal is demodulated by the PN code α. In step 90, the resultantsignal is demodulated by the channel separation function w_(Q).Concurrently, the resultant signal with respect to cos(wt) isdemodulated by the channel separation function w_(I), while theresultant signal with respect to sin(wt) is demodulated by a channelseparation function −w_(I). In step 100, the signals which weredemodulated by the channel separation function w_(Q) are thendemodulated by the channel code of a user g_(i) and the channel codep_(c).

In step 110, an integration and dump function is performed upon theresultant output signal to obtain the demodulated data signal containingthe interference. Concurrently, an integration and dump function is alsoperformed to obtain the demodulated non-orthogonal pilot signal. In step120, the demodulated non-orthogonal pilot signal is subjected to anadditional integration and dump function to remove interference from theoriginally transmitted non-orthogonal pilot signal P.

In step 130, the demodulated non-orthogonal pilot signal is modulated bythe channel code p_(c). In step 140, the modulated pilot signal ismodulated by the group user channel code g_(i). In step 150, anintegration of the signal is performed to yield first and secondinterference terms. In step 160, the signal modulated by the userchannel code g_(i) is additionally modulated by the w_(I) channelseparation function. In step 170, the resultant signal is then modulatedby the channel separation function w_(Q). In step 180, an integration ofthe resultant signal is performed to yield third and forth interferenceterms. In step 190, the interference terms are subtracted from thedemodulated data signal to remove the interference from the I and Qsub-band data. Finally, in step 200, a dot product calculation isperformed to recover the originally transmitted I and Q sub-band datasignals.

While the invention has been particularly shown and described withreference to a preferred embodiment thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the spirit and scope of theinvention.

1. In a subscriber unit, a method for increasing capacity comprising:receiving a non-orthogonal pilot signal and one or more orthogonalcommunication channels, the non-orthogonal pilot signal, modulated usinga pilot-channel code that is non-orthogonal to codes used to modulatethe one or more orthogonal communication channels, having a modulationeffect upon the one or more orthogonal communication channels; andremoving the modulation effect of the non-orthogonal pilot signal fromthe one or more orthogonal communication channels.
 2. The method ofclaim 1, wherein removing the modulation effect comprises usingdata-channel interference cancellation.
 3. The method of claim 2,wherein data-channel interference cancellation comprises: regeneratinginterference terms with respect to the non-orthogonal pilot signal; andusing the regenerated interference terms to substantially remove themodulation effects of the non-orthogonal pilot signal from the one ormore orthogonal communication channels.
 4. The method of claim 1,wherein the pilot-channel code uniquely identifies the generatednon-orthogonal pilot signal.
 5. The method of claim 1, wherein thenon-orthogonal pilot signal and the one or more orthogonal communicationchannels occupy substantially the same bandwidth at the same time.
 6. Asubscriber unit for increasing capacity comprising: a transceiver forreceiving a non-orthogonal pilot signal and one or more orthogonalcommunication channels, the non-orthogonal pilot signal, modulated usinga pilot-channel code that is non-orthogonal to codes used to modulatethe one or more orthogonal communication channels, having a modulationeffect upon the one or more orthogonal communication channels, andremoving the modulation effect of the non-orthogonal pilot signal fromthe one or more orthogonal communication channels.
 7. The subscriberunit of claim 6, wherein said transceiver comprises a pilot despreaderfor removing the modulation effect comprises means for regeneratinginterference terms with respect to the non-orthogonal pilot signal; andan interference canceller for using the regenerated interference termsto substantially remove the modulation effects of the non-orthogonalpilot signal from the one or more orthogonal communication channels. 8.The subscriber unit of claim 6, wherein the pilot-channel code uniquelyidentifies the generated non-orthogonal pilot signal.
 9. The subscriberunit of claim 6, wherein the non-orthogonal pilot signal and the one ormore orthogonal communication channels occupy substantially the samebandwidth at the same time.